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1 lt3436 3436f applicatio s u features typical applicatio u descriptio u 3a, 800khz step-up switching regulator n constant 800khz switching frequency n wide operating voltage range: 3v to 25v n high efficiency 0.1 w /3a switch n 1.2v feedback reference voltage n 2% overall output voltage tolerance n uses low profile surface mount external components n low shutdown current: 11 m a n synchronizable from 1mhz to 1.4mhz n current-mode control n constant maximum switch current rating at all duty cycles* n available in a small thermally enhanced tssop-16 package the lt ? 3436 is an 800khz monolithic boost switching regulator. a high efficiency 3a, 0.1 w switch is included on the die together with all the control circuitry required to complete a high frequency, current-mode switching regu- lator. current-mode control provides fast transient re- sponse and excellent loop stability. new design techniques achieve high efficiency at high switching frequencies over a wide operating range. a low dropout internal regulator maintains consistent perfor- mance over a wide range of inputs from 24v systems to li- ion batteries. an operating supply current of 1ma main- tains high efficiency, especially at lower output currents. shutdown reduces quiescent current to 11 m a. maximum switch current remains constant at all duty cycles. syn- chronization capability allows an external logic level signal to increase the internal oscillator from 1mhz to 1.4mhz. full cycle-by-cycle switch current limit protection and ther- mal shutdown are provided. high frequency operation al- lows the reduction of input and output filtering components and permits the use of tiny chip inductors. the lt3436 is available in an exposed pad, 16-pin tssop package. n dsl modems n portable computers n battery-powered systems n distributed power , ltc and lt are registered trademarks of linear technology corporation. efficiency vs load current 5v to 12v boost converter *patent numbers 6,535,042; 6,611,131; 6,498,466 lt3436 v in output 12v 0.9a ? input 5v 3436 ta01 10nf 470pf 4.7k 10k 1% 90.9k b220a 22 m f ceramic 4.7 m f ceramic v sw fb shdn open or high = on gnd v c sync ? maximum output current is subject to thermal derating. 3.9 m h load current (a) efficiency (%) 3436 ta01b 90 85 80 75 70 65 60 0 0.2 0.4 0.5 0.1 0.3 0.6 0.7 0.8 v in = 5v v out = 12v
2 lt3436 3436f parameter condition min typ max units recommended operating voltage l 325v maximum switch current limit l 346 a oscillator frequency 3.3v < v in < 25v l 640 800 960 khz switch on voltage drop i sw = 3a l 330 550 mv v in undervoltage lockout (note 3) l 2.47 2.6 2.73 v v in supply current i sw = 0a l 1 1.3 ma v in supply current/i sw i sw = 3a 15 ma/a shutdown supply current v shdn = 0v, v in = 25v, v sw = 25v 11 25 m a l 45 m a feedback voltage 3v < v in < 25v, 0.4v < v c < 0.9v 1.182 1.2 1.218 v l 1.176 1.224 v fb input current l 0 C 0.2 C 0.4 m a fb to v c voltage gain 0.4v < v c < 0.9v 150 350 fb to v c transconductance d i vc = 10 m a l 500 850 1300 m mho v c pin source current v fb = 1v l C 85 C 120 C 165 m a v c pin sink current v fb = 1.4v l 70 110 165 m a v c pin to switch current transconductance 4.8 a/v v c pin minimum switching threshold duty cycle = 0% 0.3 v v c pin 3a i sw threshold 0.9 v maximum switch duty cycle v c = 1.2v, i sw = 350ma 85 90 % v c = 1.2v, i sw = 1a l 80 % shdn threshold voltage l 1.28 1.35 1.42 v shdn input current (shutting down) shdn = 60mv above threshold l C7 C10 C13 m a shdn threshold current hysteresis shdn = 100mv below threshold 4 7 10 m a sync threshold voltage 1.5 2.2 v sync input frequency 1 1.4 mhz sync pin resistance i sync = 1ma 20 k w absolute m axi m u m ratings w ww u electrical characteristics package/order i n for m atio n w uu input voltage .......................................................... 25v switch voltage ......................................................... 35v shdn pin ............................................................... 25v fb pin current ....................................................... 1ma sync pin current .................................................. 1ma operating junction temperature range (note 2) lt3436e .......................................... C 40 c to 125 c storage temperature range ................ C 65 c to 150 c lead temperature (soldering, 10 sec)................. 300 c order part number (note 1) t jmax = 125 c, q ja = 45 c/w, q jc(pad) = 10 c/w consult ltc marketing for parts specified with wider operating temperature ranges. 3436efe fe part marking lt3436efe the l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at t a = 25 c. v in = 15v, v c = 0.8v, shdn, sync and switch open unless otherwise noted. fe package 16-lead plastic tssop exposed pad is gnd (pin 17), must be soldered to pcb 1 2 3 4 5 6 7 8 top view 16 15 14 13 12 11 10 9 gnd v in sw sw gnd gnd nc gnd gnd nc sync v c fb shdn nc gnd 17
3 lt3436 3436f note 1: absolute maximum ratings are those values beyond which the life of a device may be impaired. note 2: the lt3436e is guaranteed to meet performance specifications from 0 c to 125 c. specifications over the C 40 c to 125 c operating junction temperature range are assured by design, characterization and correlation with statistical process controls. note 3: minimum input voltage is defined as the voltage where the internal regulator enters lockout. actual minimum input voltage to maintain a regulated output will depend on output voltage and load current. see applications information. typical perfor m a n ce characteristics uw fb voltage switch on voltage drop oscillator frequency shdn input current shdn threshold shdn supply current electrical characteristics temperature ( c) ?0 fb voltage (v) 100 3436 g01 050 1.220 1.215 1.210 1.205 1.200 1.195 1.190 1.185 1.180 25 25 75 125 switch current (a) 0 switch voltage (mv) 0.5 1.0 1.5 2.0 3436 g02 2.5 500 450 400 350 300 250 200 150 100 50 0 3.0 temperature ( c) ?0 oscillator frequency (khz) 100 3436 g03 050 920 890 860 830 800 770 740 710 680 25 25 75 125 shdn threshold (v) 1.40 1.38 1.36 1.34 1.32 1.30 3436 g04 input voltage (v) v in current ( a) 14 12 10 8 6 4 2 0 3436 g05 0 5 10 15 20 25 30 temperature ( c) ?0 shdn input current ( a) ?2 ?0 ? ? ? ? 0 25 75 3436 g06 ?5 0 50 100 125 t a = 125 c t a = 25 c t a = 40 c temperature ( c) 50 100 050 25 25 75 125 t a = 25 c shdn = 0v shutting down starting up
4 lt3436 3436f typical perfor a ce characteristics uw gnd (pins 1, 5, 6, 8, 9, 16, 17): short gnd pins 1, 5, 6,8, 9, 16 and the exposed pad (pin 17) on the pcb. the gnd is the reference for the regulated output, so load regulation will suffer if the ground end of the load is not at the same voltage as the gnd of the ic. this condition occurs when the load current flows through the metal path between the gnd pins and the load ground point. keep the ground path short between the gnd pins and the load and use a ground plane when possible. keep the path between the input bypass and the gnd pins short. the exposed pad should be attached to a large copper area to improve thermal performance. sw (pins 3, 4): the switch pin is the collector of the on- chip power npn switch and has large currents flowing through it. keep the traces to the switching components as short as possible to minimize radiation and voltage spikes. v in (pin 2): this pin powers the internal circuitry and internal regulator. keep the external bypass capacitor close to this pin. shdn (pin 11): the shutdown pin is used to turn off the regulator and to reduce input drain current to a few microamperes. the 1.35v threshold can function as an accurate undervoltage lockout (uvlo), preventing the regulator from operating until the input voltage has reached a predetermined level. float or pull high to put the regula- tor in the operating mode. fb (pin 12): the feedback pin is used to set output voltage using an external voltage divider that generates 1.2v at the pin with the desired output voltage. if required, the current limit can be reduced during start up when the fb pin is below 0.5v (see the current limit foldback graph in the typical performance characteristics section). an imped- ance of less than 5k w at the fb pin is needed for this feature to operate. v c (pin 13): the v c pin is the output of the error amplifier and the input of the peak switch current comparator. it is normally used for frequency compensation, but can do double duty as a current clamp or control loop override. this pin sits at about 0.3v for very light loads and 0.9v at maximum load. sync (pin 14): the sync pin is used to synchronize the internal oscillator to an external signal. it is directly logic compatible and can be driven with any signal between 20% and 80% duty cycle. the synchronizing range is equal to initial operating frequency, up to 1.4mhz. see synchronization section in applications information for details. when not in use, this pin should be grounded. shdn supply current input supply current current limit foldback pi n fu n ctio n s uuu shdn voltage (v) 0 v in current ( a) 300 250 200 150 100 50 0 0.6 1.0 3436 g07 0.2 0.4 0.8 1.2 1.4 input voltage (v) 0 v in current ( a) 1200 1000 800 600 400 200 0 5 10 15 20 3436 g08 25 30 feedback voltage (v) 0 0.2 switch peak current (a) 0.4 0.8 0.6 1.0 1.2 3436 g09 4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0 fb input current ( a) 40 30 20 10 0 t a = 25 c v in = 15v t a = 25 c t a = 25 c minimum input voltage switch current
5 lt3436 3436f amplifier commands current to be delivered to the output rather than voltage. a voltage fed system will have low phase shift up to the resonant frequency of the inductor and output capacitor, then an abrupt 180 shift will occur. the current fed system will have 90 phase shift at a much lower frequency, but will not have the additional 90 shift until well beyond the lc resonant frequency. this makes it much easier to frequency compensate the feedback loop and also gives much quicker transient response. a comparator connected to the shutdown pin disables the internal regulator, reducing supply current. the lt3436 is a constant frequency, current-mode boost converter. this means that there is an internal clock and two feedback loops that control the duty cycle of the power switch. in addition to the normal error amplifier, there is a current sense amplifier that monitors switch current on a cycle-by-cycle basis. a switch cycle starts with an oscilla- tor pulse which sets the r s flip-flop to turn the switch on. when switch current reaches a level set by the inverting input of the comparator, the flip-flop is reset and the switch turns off. output voltage control is obtained by using the output of the error amplifier to set the switch current trip point. this technique means that the error figure 1. block diagram block diagra m w C + C + s input 2.5v bias regulator 800khz oscillator sw fb v c gnd 3436 f01 slope comp 0.005 w internal v cc current sense amplifier voltage gain = 40 sync shdn shutdown comparator current comparator error amplifier g m = 850 mho r s flip-flop driver circuitry s r 0.3v q1 power switch 1.2v C + C + 1.35v 3 a 7 a
6 lt3436 3436f applicatio n s i n for m atio n wu u u fb resistor network the suggested resistance (r2) from fb to ground is 10k 1%. this reduces the contribution of fb input bias current to output voltage to less than 0.2%. the formula for the resistor (r1) from v out to fb is: r rv ra out 1 212 12 202 = - () -m . .(.) figure 2. feedback network output capacitor step-up regulators supply current to the output in pulses. the rise and fall times of these pulses are very fast. the output capacitor is required to reduce the voltage ripple this causes. the rms ripple current can be calculated from: iivvv ripple rms out out in in () =- () / the lt3436 will operate with both ceramic and tantalum output capacitors. ceramic capacitors are generally cho- sen for their small size, very low esr (effective series resistance), and good high frequency operation, reducing output ripple voltage. their low esr removes a useful zero in the loop frequency response, common to tantalum capacitors. to compensate for this, the v c loop compen- sation pole frequency must typically be reduced by a factor of 10. typical ceramic output capacitors are in the 4.7 m f C + 1.2v v sw v c gnd 3436 f02 r1 r2 10k output error amplifier fb lt3436 + to 22 m f range. since the absolute value of capacitance defines the pole frequency of the output stage, an x7r or x5r type ceramic, which have good temperature stability, is recommended. tantalum capacitors are usually chosen for their bulk capacitance properties, useful in high transient load appli- cations. esr rather than absolute value defines output ripple at 800khz. values in the 22 m f to 100 m f range are generally needed to minimize esr and meet ripple current ratings. care should be taken to ensure the ripple ratings are not exceeded. table 1. surface mount solid tantalum capacitor esr and ripple current e case size esr (max, w ) ripple current (a) avx tps, sprague 593d 0.1 to 0.3 0.7 to 1.1 d case size avx tps, sprague 593d 0.1 to 0.3 0.7 to 1.1 c case size avx tps 0.2 (typ) 0.5 (typ) input capacitor unlike the output capacitor, rms ripple current in the input capacitor is normally low enough that ripple current rating is not an issue. the current waveform is triangular, with an rms value given by: i vv v lfv ripple rms in out in out () = () - () ()()( ) 029 . at higher switching frequency, the energy storage require- ment of the input capacitor is reduced so values in the range of 2.2 m f to 10 m f are suitable for most applications. y5v or similar type ceramics can be used since the absolute value of capacitance is less important and has no significant effect on loop stability. if operation is required close to the minimum input voltage required by either the output or the lt3436, a larger value may be necessary. this is to prevent excessive ripple causing dips below the minimum operating voltage resulting in erratic operation.
7 lt3436 3436f applicatio n s i n for m atio n wu u u inductor choice and maximum output current when choosing an inductor, there are 2 conditions that limit the minimum inductance; required output current, and avoidance of subharmonic oscillation. the maximum output current for the lt3436 in a standard boost con- verter configuration with an infinitely large inductor is: ia v v out max in out () = 3 h where h = converter efficiency (typically 0.87 at high current). as the value of inductance is reduced, ripple current increases and i out(max) is reduced. the minimum induc- tance for a required output current is given by: l vv v vf vi v min in out in out out out in = ? ? ? ? () () ()() 23 h the second condition, avoidance of subharmonic oscilla- tion, must be met if the operating duty cycle is greater than 50%. the slope compensation circuit within the lt3436 prevents subharmonic oscillation for inductor ripple cur- rents of up to 1.4a p-p , defining the minimum inductor value to be: l vv v vf min in out in out = () .() 14 these conditions define the absolute minimum induc- tance. however, it is generally recommended that to prevent excessive output noise, and difficulty in obtaining stability, the ripple current is no more than 40% of the average inductor current. since inductor ripple is: i vv v vlf p p ripple in out in out - = () ()() the recommended minimum inductance is: l vv v vif min in out in out out = ()( ) . ( ) ( )( ) 2 2 04 the inductor value may need further adjustment for other factors such as output voltage ripple and filtering require- ments. remember also, inductance can drop significantly with dc current and manufacturing tolerance. the inductor must have a rating greater than its peak operating current to prevent saturation resulting in effi- ciency loss. peak inductor current is given by: i vi v vv v vlf lpeak out out in in out in out =+ - ()() () ()() h 2 also, consideration should be given to the dc resistance of the inductor. inductor resistance contributes directly to the efficiency losses in the overall converter. suitable inductors are available from coilcraft, coiltronics, dale, sumida, toko, murata, panasonic and other manu- factures. table 2 part value i sat(dc) dcr height number ( m h) (amps) ( w ) (mm) coilcraft do1608c-222 2.2 2.4 0.07 2.9 sumida cdrh3d16-1r5 1.5 1.6 0.043 1.8 cdrh4d18-1r0 1.0 1.7 0.035 2.0 cdc5d23-2r2 2.2 2.2 0.03 2.5 cr43-1r4 1.4 2.5 0.056 3.5 cdrh5d28-2r6 2.6 2.6 0.013 3.0 cdrh6d38-3r3 3.3 3.5 0.02 4.0 cdrh6d28-3r0 3.0 3.0 0.024 3.0 toko (d62f)847fy-2r4m 2.4 2.5 0.037 2.7 (d73lf)817fy-2r2m 2.2 2.7 0.03 3.0
8 lt3436 3436f applicatio n s i n for m atio n wu u u shutdown pin can be used. the threshold voltage of the shutdown pin comparator is 1.35v. a 3 m a internal current source defaults the open pin condition to be operating (see typical performance graphs). current hysteresis is added above the shdn threshold. this can be used to set voltage hysteresis of the uvlo using the following: r vv a r v vv r a hl h 1 7 2 135 135 1 3 = - m = - () +m . . v h e turn-on threshold v l e turn-off threshold example: switching should not start until the input is above 4.75v and is to stop if the input falls below 3.75v. v h = 4.75v v l = 3.75v r vv a k r v vv k a k 1 475 375 7 143 2 135 475 135 143 3 50 4 = - m = = - () +m = .. . .. . keep the connections from the resistors to the shdn pin short and make sure that the interplane or surface capaci- tance to the switching nodes are minimized. if high resis- tor values are used, the shdn pin should be bypassed with a 1nf capacitor to prevent coupling problems from the switch node. catch diode the suggested catch diode (d1) is a b220a schottky. it is rated at 2a average forward current and 20v reverse voltage. typical forward voltage is 0.5v at 2a. the diode conducts current only during switch off time. peak reverse voltage is equal to regulator output voltage. average forward current in normal operation is equal to output current. shutdown and undervoltage lockout figure 4 shows how to add undervoltage lockout (uvlo) to the lt3436. typically, uvlo is used in situations where the input supply is current limited , or has a relatively high source resistance. a switching regulator draws constant power from the source, so source current increases as source voltage drops. this looks like a negative resistance load to the source and can cause the source to current limit or latch low under low source voltage conditions. uvlo prevents the regulator from operating at source voltages where these problems might occur. figure 4. undervoltage lockout 1.35v gnd input r1 3436 f04 shdn v cc in lt3436 3 a r2 c1 7 a an internal comparator will force the part into shutdown below the minimum v in of 2.6v. this feature can be used to prevent excessive discharge of battery-operated sys- tems. if an adjustable uvlo threshold is required, the
9 lt3436 3436f synchronization the sync pin, is used to synchronize the internal oscilla- tor to an external signal. the sync input must pass from a logic level low, through the maximum synchronization threshold with a duty cycle between 20% and 80%. the input can be driven directly from a logic level output. the synchronizing range is equal to initial operating frequency up to 1.4mhz. this means that minimum practical sync frequency is equal to the worst-case high self-oscillating frequency (960khz), not the typical operating frequency of 800khz. caution should be used when synchronizing above 1.1mhz because at higher sync frequencies the amplitude of the internal slope compensation used to prevent subharmonic switching is reduced. higher induc- tor values will tend to eliminate this problem. see fre- quency compensation section for a discussion of an entirely different cause of subharmonic switching before assuming that the cause is insufficient slope compensa- tion. application note 19 has more details on the theory of slope compensation. layout considerations as with all high frequency switchers, when considering layout, care must be taken to achieve optimal electrical, thermal and noise performance. for maximum efficiency, switch rise and fall times are typically in the nanosecond range. to prevent noise both radiated and conducted, the applicatio n s i n for m atio n wu u u high speed switching current path, shown in figure 5, must be kept as short as possible. this is implemented in the suggested layout of figure 6. shortening this path will also reduce the parasitic trace inductance of approxi- mately 25nh/inch. at switch off, this parasitic inductance produces a flyback spike across the lt3436 switch. when operating at higher currents and output voltages, with poor layout, this spike can generate voltages across the lt3436 that may exceed its absolute maximum rating. a ground plane should always be used under the switcher circuitry to prevent interplane coupling and overall noise. the v c and fb components should be kept as far away as possible from the switch node. the lt3436 pinout has been designed to aid in this. the ground for these compo- nents should be separated from the switch current path. failure to do so will result in poor stability or subharmonic like oscillation. board layout also has a significant effect on thermal resistance. the exposed pad is the copper plate that runs under the lt3436 die. this is the best thermal path for heat out of the package. soldering the pad onto the board will reduce die temperature and increase the power capability of the lt3436. provide as much copper area as possible around this pad. adding multiple solder filled feedthroughs under and around the pad to the ground plane will also help. similar treatment to the catch diode and inductor terminations will reduce any additional heating effects. figure 5. high speed switching path 3436 f05 v out l1 sw gnd lt3436 d1 c1 c3 v in high frequency switching path load
10 lt3436 3436f figure 6. typical application and suggested layout (topside only shown) v out input gnd c3 c1 r2 r1 l1 d1 kelvin sense v out minimize lt3436, c1, d1 loop place feedthroughs around ground pin for good thermal conductivity c4 u1 solder exposed ground pad to board keep fb and v c components away from high frequency, high input components c2 lt3436 v in output 12v 0.8a ? input 5v c2 10nf c4 470pf r3 4.7k r2 10k 1% r1 90.9k d1 b220a c1 22 f ceramic c3 4.7 f ceramic v sw fb shdn open or high = on gnd v c sync ? maximum output current is subject to thermal derating. l1 3.9 h gnd r3 applicatio n s i n for m atio n wu u u
11 lt3436 3436f applicatio n s i n for m atio n wu u u the inductor must have a rating greater than its peak operating current to prevent saturation resulting in effi- ciency loss. peak inductor current is given by: i vi v vv v vlf lpeak out out in in out in out =+ - ()() () ()() h 2 also, consideration should be given to the dc resistance of the inductor. inductor resistance contributes directly to the efficiency losses in the overall converter. thermal calculations power dissipation in the lt3436 chip comes from four sources: switch dc loss, switch ac loss, drive current, and input quiescent current. the following formulas show how to calculate each of these losses. these formulas assume continuous mode operation, so they should not be used for calculating efficiency at light load currents. dc duty cycle vv v i vi v out in out sw out out in , () ()() = - = switch loss: pdcir nivf sw sw sw sw out =+ () () () ()( )( ) 2 17 v in loss: p vi dc ma v vin in sw in =+ ()( )() () 50 1 r sw = switch resistance ( ? 0.16 w hot) example: v in = 5v, v out = 12v and i out = 0.8a total power dissipation = 0.34 + 0.31 + 0.11 + 0.005 = 0.77w thermal resistance for the lt3436 package is influenced by the presence of internal or backside planes. with a full plane under the package, thermal resistance will be about 40 c/w. to calculate die temperature, use the appropriate thermal resistance number and add in worst-case ambient temperature: t j = t a + q ja (p tot ) if a true die temperature is required, a measurement of the sync to gnd pin resistance can be used. the sync pin resistance across temperature must first be calibrated, with no device power, in an oven. the same measurement can then be used in operation to indicate the die tempera- ture. frequency compensation loop frequency compensation is performed on the output of the error amplifier (v c pin) with a series rc network. the main pole is formed by the series capacitor and the output impedance ( ? 500k w ) of the error amplifier. the pole falls in the range of 2hz to 20hz. the series resistor creates a zero at 1khz to 5khz, which improves loop stability and transient response. a second capacitor, typi- cally one-tenth the size of the main compensation capaci- tor, is sometimes used to reduce the switching frequency ripple on the v c pin. v c pin ripple is caused by output voltage ripple attenuated by the output divider and multi- plied by the error amplifier. without the second capacitor, v c pin ripple is: v c pin ripple = v ripple = output ripple (v p? ) g m = error amplifier transconductance ( 850 mho) r c = series resistor on v c pin v out = dc output voltage 1.2(v ripple )(g m )(r c ) (v out ) to prevent irregular switching, v c pin ripple should be kept below 50mv pCp . worst-case v c pin ripple occurs at maximum output load current and will also be increased if poor quality (high esr) output capacitors are used. the addition of a 150pf capacitor on the v c pin reduces switching frequency ripple to only a few millivolts. a low value for r c will also reduce v c pin ripple, but loop phase margin may be inadequate.
12 lt3436 3436f v in (v) 0 maximum load current (ma) 16 14 12 18 3436 ta02c 48 2.0 1.8 1.6 1.4 1.2 1.0 0.8 0.6 0.4 0.2 0 2 6 10 20 load current (ma) efficiency (%) 0 1.5k 1.0k 3436 ta02d 500 100 90 80 70 60 50 40 30 20 10 0 2.0k 3.3v in 5v in 12v in lt3436 v in fb v c sync gnd lt3436 ? ta02 v sw shdn d2 1n4148 d3 1n4148 c3 4.7 m f c1 4.7 m f c6 0.1 m f c2 10nf c4 470pf r3 4.7k r1 90.9k r2 10k 1% r4 1m v out 12v 0.8a c7 22 m f c5 0.1 m f off on v in 5v d1 b220a l1 3.9 h q1 si2306ds load disconnects in shutdown typical applicatio s u 3v to 20v in 5v out sepic with either two inductors or a transformer v in fb v c sync gnd gnd 3436 ta02b sw shdn c1 opt c3 10nf c4 470pf r3 2.2k r1 31.6k 1% r2 10k 1% v out 5v gnd gnd c6 opt c5 opt v in 3v to 20v d1 b220a l1 cdrh6d28-100 c1 4.7 m f x5r 25v ceramic c2 22 f x5r 10v ceramic c7 1 f, x5r, 25v ceramic l2 cdrh6d28-100 + lt3436 sync shdn option: replace l1, l2 with transformer ctx5-1a, ctx8-1a, ctx10-2a maximum load current increases with input voltage efficiency
13 lt3436 3436f lt3436 v in gnd v in ** 4v to 9v v c fb lt3436 ? ta03 v sw shdn c1 4.7 m f 20v c4 15nf c5 470pf r1 2.2k r3 10k 1% r2 31.6k 1% v out ? 5v c3 47 f 10v on off l1a* 15 h ? ? l1b* 15 h c2 4.7 f coiltronics ctx15-4 input voltage may be greater or less than output voltage d1 b220a v in 4v 5v 6v 7v 9v i out 0.84a 1.03a 1.18a 1.29a 1.50a ? max i out * ** + + 4v-9v in to 5v out sepic converter** typical applicatio s u boost converter drives luxeon iii 1a 3.6v white led with 70% efficiency v in fb v c sync gnd gnd 3436 ta03b sw shdn 0.1 f q1 q2 v out = v in + v led v out v in gnd v in 3.3v to 4.2v lt3436 ups120 l1 0.05 1% 1a constant current q1: mmbt2222a q2: fmmt3906 l1: cdrh6d28-3r0 lxhl-pw09 emitter 4.7 m f x5r 6.3v ceramic 22 f x5r 10v ceramic 1.21k 1% 4.99k 78.7k 49.9 1% 8.2k C + led on lt1783
14 lt3436 3436f lt3436 v in v c gnd fb lt3436 ? ta04 v sw shdn l1 4.7 h c1 10 m f single li-ion cell c4 47 f 10v c2 3.3nf c3 470pf r3 1.5k r2 10k 1% r1 31.6k 1% v out 5v d1 b220a on off + + + v in 2.7v 3.3v 3.6v i out 1.5a 1.86a 2.0a single li-ion cell to 5v typical applicatio s u sepic converter drives 5w lumileds luxeon v white leds at 70% efficiency v in fb v c sync gnd gnd 3436 ta04b sw shdn c4 0.1 f q1 v out v out v in gnd v in 3.6v to 17v lt3436 d1 b130a l1 l2 r7 124k q1: diodes, inc. mmbt2222a l1: cdrh6d28 10 h 1.7a l2: cdrh4d28 10 h 1a d2: lumileds lxhl-pw03 emitter or lxhl-lw6c star d2 700ma c1 4.7 m f x5r 25v ceramic c2 22 f x5r 16v ceramic c coup 2.2 f, x5r, 25v ceramic r6 4.99k 8.2k led on C + r4 1k 1% r2 0.068 1% r5 23.7k lt1783
15 lt3436 3436f package descriptio n u fe package 16-lead plastic tssop (4.4mm) (reference ltc dwg # 05-08-1663) exposed pad variation bb fe16 (bb) tssop 0204 0.09 ?0.20 (.0035 ?.0079) 0 ?8 0.25 ref 0.50 ?0.75 (.020 ?.030) 4.30 ?4.50* (.169 ?.177) 134 5 6 7 8 10 9 4.90 ?5.10* (.193 ?.201) 16 1514 13 12 11 1.10 (.0433) max 0.05 ?0.15 (.002 ?.006) 0.65 (.0256) bsc 2.94 (.116) 0.195 ?0.30 (.0077 ?.0118) typ 2 recommended solder pad layout 0.45 0.05 0.65 bsc 4.50 0.10 6.60 0.10 1.05 0.10 2.94 (.116) 3.58 (.141) 3.58 (.141) millimeters (inches) *dimensions do not include mold flash. mold flash shall not exceed 0.150mm (.006") per side note: 1. controlling dimension: millimeters 2. dimensions are in 3. drawing not to scale see note 4 4. recommended minimum pcb metal size for exposed pad attachment 6.40 (.252) bsc information furnished by linear technology corporation is believed to be accurate and reliable. however, no responsibility is assumed for its use. linear technology corporation makes no represen- tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
16 lt3436 3436f part number description comments lt1310 1.5a (i sw ), 4.5 mhz, high efficiency v in = 2.75v to 18v, v out(max) = 35v, i q = 12ma, step-up dc/dc converter with pll i sd = <1 m a, mse package lt1370/lt1370hv 6a (isw), 500khz, high efficiency v in = 2.7v to 30v, v out(max) = 35v/42v, i q = 4.5ma, step-up dc/dc converter i sd = <12 m a, dd, to220-7 packages lt1371/lt1371hv 3a (i sw ), 500khz, high efficiency v in = 2.7v to 30v, v out(max) = 35v/42v, i q = 4ma, step-up dc/dc converter i sd = <12 m a, dd,to220-7,s20 packages lt1613 550ma (i sw ), 1.4mhz, high efficiency 90% efficiency, v in = 0.9v to 10v, v out(max) = 34v, i q = 3ma, step-up dc/dc converter i sd = <1 m a, thinsot package lt1618 1.5a (i sw ), 1.25mhz, high efficiency 90% efficiency, v in = 1.6v to 18v, v out(max) = 35v, i q = 1.8ma, step-up dc/dc converter i sd = <1 m a, ms package lt1946/lt1946a 1.5a (i sw ), 1.2mhz/2.7mhz, high efficiency v in = 2.45v to 16v, v out(max) = 34v, i q = 3.2ma, step-up dc/dc converter i sd = <1 m a, ms8 package lt1961 1.5a (i sw ), 1.25mhz, high efficiency 90% efficiency, v in = 3v to 25v, v out(max) = 35v, i q = 0.9ma, step-up dc/dc converter i sd = 6 m a, ms8e package ltc3400/ltc3400b 600ma (i sw ), 1.2mhz, synchronous 92% efficiency, v in = 0.85v to 5v, v out(max) = 5v, step-up dc/dc converter i q = 19 m a/300 m a, i sd = <1 m a, thinsot package ltc3401 1a (i sw ), 3mhz, synchronous 97% efficiency, v in = 0.5v to 5v, v out(max) = 6v, i q = 38 m a, step-up dc/dc converter i sd = <1 m a, ms package ltc3402 2a (i sw ), 3mhz, synchronous 97% efficiency, v in = 0.5v to 5v, v out(max) = 6v, i q = 38 m a, step-up dc/dc converter i sd = <1 m a, ms package lt/tp 0504 1k ? printed in usa ? linear technology corporation 2003 related parts linear technology corporation 1630 mccarthy blvd., milpitas, ca 95035-7417 (408) 432-1900 l fax: (408) 434-0507 l www.linear.com laser 190 w 1% 1n4002 (all) 0.1 m f 10k v in 10 m f v c v in fb gnd 2.2 m f v in 12v to 25v 150 w mur405 l2 10 m h lt3436 l1 5 4 1 3 2 8 11 hv diodes 1800pf 10kv 0.01 m f 5kv 1800pf 10kv 47k 5w 2.2 m f 0.47 m f l1 = q1, q2 = 0.47 m f = hv diodes = laser = tbd zetex ztx849 wima 3x 0.15 m f type mkp-20 semtech-fm-50 hughes 3121h-p 10k lt3436 ? ta05 v sw q1 q2 + + + coiltronics (407) 241-7876 u typical applicatio high voltage laser power supply

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